High-frequency signal processor and wireless communication system

ABSTRACT

There is a need to reduce secondary intermodulation distortion that may occur in a reception circuit of a high-frequency signal processor and a wireless communication system having the same. In test mode, for example, a test signal generating circuit TSGEN generates a test signal RFtst at f_tx ±0.5 MHz. The test signal RFtst is input to a mixer circuit MIXrx_I (MIXrx_Q). A correction circuit block CALBK detects an IM2 component resulting from the MIXrx_I (MIXrx_Q). The CALBK varies a differential balance for the MIXrx_I (MIXrx_Q) and concurrently monitors a phase for the IM2 component resulting from MIXrx_I (MIXrx_Q). The CALBK searches for the differential balance corresponding to a transition point that allows the phase to transition by approximately 180°. The MIXrx_I (MIXrx_Q) operates in normal mode using the differential balance as a search result.

CROSS-REFERENCE TO RELATED APPLICATIONS

The disclosure of Japanese Patent Application No. 2011-239887 filed onNov. 1, 2011 including the specification, drawings and abstract isincorporated herein by reference in its entirety.

BACKGROUND

The present invention relates to a high-frequency signal processor and awireless communication system. More particularly, the invention relatesto a technology effectively applicable to a high-frequency signalprocessor having a direct conversion receiver and a wirelesscommunication system.

As described in patent document 1; for example, the direct conversionreceiver reduces secondary distortion generated in the mixer using thefollowing circuits. One circuit supplies the mixer with a test signalhaving a predetermined frequency interval. Another circuit detectssecondary distortion generated in the mixer. Still another circuitcontrols mixer parameters based on the detection result. Thisconfiguration searches for a mixer parameter to minimize the secondarydistortion in the mixer. When detecting the secondary distortion in themixer, the receiver detects an output amplitude magnitude from the mixerbased on a specific frequency causing the secondary distortion.

Patent document 2 describes the technology that allows a directconversion transmitter to reduce carrier leakage occurring in a firstmodulator (I-signal mixer circuit) and a second modulator (Q-signalmixer circuit). For example, carrier leakage may be reduced in the firstmodulator. For this purpose, the transmitter detects a phase differencebetween a local signal for the first modulator and an added outputsignal from the first modulator and the second modulator while changinga differential balance for the first modulator. The transmitter searchesfor a differential balance that allows the phase difference to reach aspecified value (90°), namely that allows the carrier leakage to remainin only the second modulator.

Patent document 1: Japanese patent laid-open No. 2004-336822

Patent document 2: Japanese patent laid-open No. 2009-212869

SUMMARY

FIG. 15 is a block diagram illustrating a concise configuration examplein a wireless communication system as a prerequisite for the presentinvention. For example, the wireless communication system is representedas a mobile telephone. As illustrated in FIG. 15, The wirelesscommunication system includes a high-frequency signal processing chip(high-frequency signal processor) RFIC′ that mainly converts a frequencybetween a baseband frequency band and a high frequency band (RadioFrequency (RF) band). The RFIC′ includes a low-noise amplifier circuitLNA, a mixer circuit MIX, and a driver circuit (variable amplifiercircuit) DRV as reception circuits. The low-noise amplifier circuit LNAamplifies a high-frequency signal received at an antenna ANT. The mixercircuit MIX is provided after the low-noise amplifier circuit LNA andconverts a high frequency band into the baseband. The RFIC′ includes adriver circuit (variable amplifier circuit) DRV as a transmissioncircuit. The DRV is provided before a high-frequency power amplifiercircuit HPA.

For example, a SAW (Surface Acoustic Wave) filter SAWrx is providedbetween LNA and MIX. The SAWrx is externally added to the chip andremoves an unnecessary frequency band other than the reception band. ASAW filter SAWtx is provided between DRV and HPA. The SAWtx isexternally added to the chip and removes an unnecessary frequency bandother than the transmission band. Recently, there is an increasingdemand for miniaturizing the wireless communication system including thehigh-frequency signal processing chip and reducing costs on the system.Therefore, the high-frequency signal processing chip needs to eliminatethe SAW filters.

If the SAW filter is eliminated, however, a leakage signal from thetransmission circuit may be superposed on a targeted baseband signal viaa secondary intermodulation distortion (IM2) in the reception circuit.FIGS. 16A through 16C are explanatory diagrams illustrating a problem onthe high-frequency signal processor as a prerequisite for the presentinvention. For example, the FDD (Frequency Division Duplex) systemincludes W-CDMA (Wideband Code Division Multiple Access) and LTE (LongTerm Evolution). The high-frequency power amplifier circuit HPAillustrated in FIG. 15 may output a transmission signal having a largeamount of power. The transmission signal may then leak to the receptioncircuit via a duplexer DPX. FIG. 16A illustrates an output frequencyspectrum in the reception circuit in this situation.

As illustrated in FIG. 16A, the LNA is supplied with a high-frequencysignal (targeted wave signal) RFrx within the reception band and atransmission leakage signal RFtx_L from the HPA. The LNA amplifies thesesignals and outputs a result. The RFrx is received at the antenna ANT.The RFrx has a specified signal band (2·f_BB) at several megahertzes.Similarly, the RFtx_L also has a specified signal band. In the example,the signal band for the RFtx_L is represented as two frequency spectrahaving a specified frequency interval (f_int) such as 1 MHz. The RFtx_Lis sufficiently suppressed at the input and output of the LNA if theSAWtx and the SAWrx are available. The RFtx_L is not suppressed and issupplied to the mixer circuit MIX if the SAWtx and the SAWrx areunavailable.

FIG. 16B illustrates a local signal (locally-generated signal) LOrxconfigured for a specified channel frequency in the reception band. Themixer circuit MIX multiplies the LOrx by the targeted wave signal RFrx.As illustrated in FIG. 16C, the MIX directly down-coverts the RFrx intothe baseband frequency band (f_BB) as frequency conversion and outputs aresulting reception baseband signal BBrx. Here, the mixer circuit MIXmay generate an IM2 component because of a device mismatch. The MIXgenerates an interfering wave at the frequency of f_int if the MIXgenerates the IM2 component because of the transmission leakage signalRFtx_L. As illustrated in FIG. 16C, the interfering wave (f_int) issuperposed on the frequency band (f_BB) for a reception baseband signalBBrx. Hence, a normal reception operation is difficult.

To solve the problem of the IM2, the technology described in patentdocument 1 may be used. The system used in patent document 1 observesthe amplitude level magnitude of the IM2 component and searches for acorrection parameter that minimizes the magnitude. For this purpose, thesystem searches the I-signal mixer circuit and the Q-signal mixercircuit independently of each other. Alternatively, the system searchesone of the mixer circuits. The system applies the search result to theother mixer circuit, assuming that the other mixer circuit yields thesame search result. Hence, the following problems may arise.

(1) The amplitude level of the IM2 component is very small near aposition (optimum correction point) where the IM2 is minimized. Thedetection itself may be difficult. If the amplitude level is detected,optimum correction points may vary to some extent depending on thedetection accuracy for the amplitude level. It may be difficult tosettle highly accurately the optimum correction point or highlyaccurately search for the minimum IM2 point and reduce the IM2. (2)Correcting the I-signal mixer circuit changes the optimum correctionpoint for the Q-signal mixer circuit. By contrast, correcting theQ-signal mixer circuit changes the optimum correction point for theI-signal mixer circuit. This IQ interference problem cannot be solved.

(3) Searching for the minimum IM2 point (optimum correction point) maybe time-consuming. A search for the minimum point for the IM2 amplitudelevel may analogize with a search for the local minimum point of aU-shaped or V-shaped curve. For example, an actual, possible searchmethod aims at a minimum point by roughly estimating correctionvariables over an entire variable range. The method then varies thevicinity of the aimed minimum point. However, this search method mayrequire many processing steps. In addition, the method needs torepeatedly correct the I-signal mixer circuit and the Q-signal mixercircuit if the IQ interference problem (2) above needs to be solved. Asa result, the search time may increase drastically.

The present invention has been made in consideration of the foregoing.The invention aims at reducing secondary intermodulation distortion thatmay result from a reception circuit in a high-frequency signal processorand a wireless communication system having the same. These and otherobjects and novel features of the invention may be readily ascertainedby referring to the following description and appended drawings.

The following summarizes representative embodiments of the inventiondisclosed in this application.

A high-frequency signal processor according to an embodiment of theinvention is provided with a first operation mode and a second operationmode, and includes a test signal generating circuit, a first switch, amixer circuit, a phase detection portion, and a control portion. Thetest signal generating circuit generates a test signal having a firstfrequency component and a second frequency component. The first switch(SWr) transmits a signal received as a first signal at an antenna in thefirst operation mode and transmits the test signal as the first signalin the second operation mode. The mixer circuit (MIXrx_I or MIXrx_Q)includes a differential circuit capable of correcting a differentialbalance within a specified variable range and down-converts the firstsignal to a second signal having a frequency band lower than the firstsignal. The phase detection portion (PHDET) extracts a third signal fromthe second signal in the second operation mode and detects a phase forthe third signal. The third signal has a frequency component equivalentto a difference between the first frequency component and the secondfrequency component. The control portion (DGCTL) changes thedifferential balance for the mixer circuit according to a detectionresult from the phase detection portion. The mixer circuit operates inthe first operation mode while the differential balance is set to afirst correction value within a variable range. In the second operationmode, the control portion varies the differential balance andconcurrently searches for a transition point allowing a phase for thethird signal to transition by approximately 180° before and aftervarying the differential balance within a minimum fluctuation range. Thecontrol portion supplies the mixer circuit with the first correctionvalue, namely, the differential balance corresponding to the transitionpoint.

According to the above-mentioned configuration, correcting thedifferential balance for the mixer circuit can reduce a secondaryintermodulation distortion (IM2) component resulting from the mixercircuit. The high-frequency signal processor searches for an optimumcorrection value for the differential balance while monitoring phaseinformation about the IM2 component output from the mixer circuit. Thisenables to facilitate the correction, provide highly accuratecorrection, and shorten the correction time. Particularly, a binarysearch for the differential balance can moreover shorten the correctiontime.

An effect of the representative embodiment discussed in the presentinvention is briefly summarized as being capable of reducing secondaryintermodulation distortion that may occur in a reception circuit of ahigh-frequency signal processor and a wireless communication systemhaving the same.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a block diagram illustrating a concise configuration exampleof major parts of the wireless communication system according to a firstembodiment of the invention;

FIG. 2 is a circuit block diagram illustrating a detailed configurationexample of major parts of a high-frequency signal processor in thewireless communication system illustrated in FIG. 1;

FIG. 3A is an explanatory diagram illustrating an example of IM2characteristics in the high-frequency signal processor illustrated inFIG. 2;

FIG. 3B is a conceptual diagram illustrating an example ofcharacteristic mechanism illustrated in FIG. 3A;

FIG. 4 is a circuit block diagram illustrating a more detailedconfiguration example of a test signal generating circuit and acorrection circuit block in the high-frequency signal processorillustrated in FIG. 2;

FIG. 5 is a circuit diagram illustrating a configuration example of areception mixer circuit in the high-frequency signal processorillustrated in FIG. 2;

FIG. 6 is a conceptual diagram illustrating an example method foradjusting differential balance (IM2 correction parameter) in the mixercircuit illustrated in FIG. 5;

FIG. 7 is a conceptual diagram illustrating another example method foradjusting differential balance (IM2 correction parameter) in the mixercircuit illustrated in FIG. 5;

FIG. 8 is a conceptual diagram illustrating still another example methodfor adjusting differential balance (IM2 correction parameter) in themixer circuit illustrated in FIG. 5;

FIG. 9 is a flowchart illustrating an example method for the correctioncircuit block in the high-frequency signal processor illustrated in FIG.2 to search for an optimum correction point;

FIG. 10 supplements an example of actual operation according to theflowchart illustrated in FIG. 9;

FIGS. 11A and 11B illustrate an example problem in a high-frequencysignal processor according to a second embodiment of the invention;

FIG. 12 is an explanatory diagram illustrating an example method for thehigh-frequency signal processor according to the second embodiment ofthe invention to search for an optimum correction point;

FIGS. 13A and 13B supplement FIG. 12;

FIG. 14 is a flowchart illustrating an example method for ahigh-frequency signal processor according to a third embodiment of theinvention to search for an optimum correction point;

FIG. 15 is a block diagram illustrating a concise configuration exampleof a wireless communication system as a prerequisite for the presentinvention; and

FIGS. 16A through 11C illustrate an example problem in a high-frequencysignal processor as a prerequisite for the present invention.

DETAILED DESCRIPTION

The following description is divided into sections or embodiments asneeded. These are related to each other unless explicitly statedotherwise. One section or embodiment may represent a modification,details, or supplementary description for all or part of the others. Theembodiments may refer to the number of elements including the number ofitems, numeric values, quantities, and ranges. The embodiments are notlimited to specific values unless explicitly stated otherwise or unlessthe embodiment or embodiments are unquestionably limited to specificvalues in principle. For example, the embodiment or embodiments may begreater, smaller than, or equal to a specific value.

Constituent elements (including process steps) of the embodiments arenot necessarily required unless explicitly stated otherwise or unlessthe constituent elements are unquestionably required in principle.Similarly, the embodiments may refer to shapes or positionalrelationship among the constituent elements. The embodiments include anequivalent substantially similar or approximate to the shapes unlessexplicitly stated otherwise or unless the equivalent is unquestionablyunavailable in principle. The same applies to the above-mentionednumeric values and ranges.

A circuit element configures each function block according to theembodiments. The circuit element is formed on a semiconductor substratemade of single-crystal silicon according to a known integrated-circuittechnology for CMOS (complementary metal-oxide semiconductor), thoughnot limited thereto. The embodiments use MOSFET (Metal OxideSemiconductor Field Effect Transistor) (abbreviated to a MOS transistor)as an example of MISFET (Metal Insulator Semiconductor Field EffectTransistor). Gate insulator films may include a non-oxidizing film.

Embodiments of the present invention will be described in further detailwith reference to the accompanying drawings. Throughout the drawings forillustrating the embodiments, the same members are generally designatedby the same reference numerals and a repetitive description is omittedfor simplicity.

First Embodiment

Entire configuration of a wireless communication system

FIG. 1 is a block diagram illustrating a concise configuration exampleof major parts of the wireless communication system according to thefirst embodiment of the invention. The wireless communication systemillustrated in FIG. 1 typically represents but is not limited to amobile telephone system for W-CDMA (Wideband Code Division MultipleAccess) and LTE (Long Term Evolution). The wireless communication systemillustrated in FIG. 1 includes a high-frequency signal processor chip(high-frequency signal processor) RFIC, a high-frequency power amplifiercircuit HPA, an isolator ISO, a duplexer DPX, and an antenna ANT. TheRFIC includes one semiconductor chip, for example. The RFIC includes alow-noise amplifier circuit LNA and a reception mixer circuit MIXrx asreception circuits. The RFIC includes a driver circuit (variableamplifier circuit) DRV and a transmission mixer circuit MIXtx astransmission circuits. The RFIC includes a back-end circuit BE as atransmission/reception circuit. The BE includes a baseband processorsuch as a CPU (Central Processing Unit) or an application processor, forexample.

During transmission, the transmission mixer circuit MIXtx in the RFICup-converts (frequency conversion) a transmission baseband signal fromthe BE using a local signal (locally-generated signal or carrier signal)LOtx having a specified frequency (a specific frequency in thetransmission frequency band). The DRV linearly amplifies an outputsignal from the MIXtx at a specified gain and outputs the signal to theHPA. The HPA is configured as one semiconductor chip, for example. TheHPA is provided with an HBT (Heterojunction Bipolar Transistor) usingcomposite semiconductor, though not limited thereto. The HPA amplifiesthe power of an output signal from the DRV. The HPA outputs theamplified high-frequency signal RFtx to the DPX via the ISO. The ISOpasses through the signal output from the HPA to the DPX and blocks areverse signal.

The DPX separates a transmission frequency band from a receptionfrequency band. Specifically, the DPX selects a specified transmissionfrequency band from the high-frequency signal RFtx output via the ISO.The DPX transmits the selected transmission frequency band as atransmission power signal TX to the ANT. The DPX selects a specifiedreception frequency band from a reception power signal RX received atthe ANT. The DPX transmits the selected reception frequency band as ahigh-frequency signal RFrx to the LNA in the RFIC. The LNA amplifies thehigh-frequency signal RFrx from the DPX and outputs it to the receptionmixer circuit MIXrx. The MIXrx directly down-converts (frequencyconversion) the output signal from the LNA to the baseband frequencyband using the local signal (locally-generated signal or carrier signal)LOrx having a specified frequency (a specific frequency in the receptionfrequency band). The MIXrx outputs a conversion result as the receptionbaseband signal BBrx to the back-end circuit BE. The BE receives theBBrx and performs a specified baseband process.

The RFIC, the HPA, the ISO, and the DPX may be mounted as independentparts on the same wiring substrate. Alternatively, the HPA, the ISO, andthe DPX may be mounted on one module wiring substrate. The module wiringsubstrate and the RFIC may be mounted on the same wiring substrate. Thewiring substrate and the module wiring substrate are typically made ofceramics. However, the invention is not limited thereto.

The wireless communication system illustrated in FIG. 1 ischaracteristically configured to remove the SAW filters SAWtx and SAWrxfrom the configuration example illustrated in FIG. 15. This canminiaturize the wireless communication system and reduce its cost.However, removing the SAW filters causes the problem described withreference to FIG. 16 in the FDD-based wireless communication system thatperforms transmission and reception in the same period. Actually, thehigh-frequency signal RFtx from the HPA may be supplied as atransmission leakage signal (RFtx_L) to the input side of the LNA viathe DPX. Generally, a differential MIXrx is prone to a variation in thedifferential pair and therefore causes secondary intermodulationdistortion (IM2) in response to the transmission leakage signal. As aresult, an interfering wave is superposed on the baseband frequencyband.

The interfering wave due to IM2 may result from the normalhigh-frequency signal (targeted wave signal) RFrx as well as thetransmission leakage signal. According to circumstances, this problemmay occur on the TDD (Time Division Duplex) system such as GSM (GlobalSystem for Mobile Communications), registered trademark, in addition tothe FDD system. Normally, however, the RFtx power level is much higherthan the RFrx power level. The transmission leakage signal is morecritical. The use of the high-frequency signal processing chip(high-frequency signal processor) according to the embodiment, to bedescribed later, advantageously solves the IM2 problem.

Configuration of Major Parts of the High-Frequency Signal Processor

FIG. 2 is a circuit block diagram illustrating a detailed configurationexample of major parts of the high-frequency signal processor in thewireless communication system illustrated in FIG. 1. FIG. 2 illustratesa detailed configuration example around the reception circuit in ahigh-frequency signal processor RFIC illustrated in FIG. 1. The RFICillustrated in FIG. 2 includes a low-noise amplifier circuit LNA, alocal signal generating circuit LOGEN, mixer circuits MIXrx_I andMIXrx_Q, filter circuits FLTi and FLTq, variable amplifier circuits PGAiand PGAq, analog-digital converter circuits ADCi and ADCq, and aback-end circuit BE. In addition to these circuits, the RFIC illustratedin FIG. 2 characteristically includes switches SWr, SWi, and SWq, a testsignal generating circuit TSGEN, and a correction circuit block CALBK.

The SWr selects one of output signals from the LNA and the TSGEN andoutputs the selected signal to the MIXrx_I and the MIXrx_Q. The MIXrx_Iuses a local signal LOrx_I from the LOGEN to down-convert the outputsignal from the SWr to the baseband frequency band. The MIXrx_Q uses alocal signal LOrx_Q from the LOGEN to down-convert the output signalfrom the SWr to the baseband frequency band. The signals LOrx_I andLOrx_Q are orthogonal to each other at a 90° phase. The MIXrx_I and theMIXrx_Q perform direct conversion and quadrature demodulation. Thoughnot illustrated, the MIXrx_I and the MIXrx_Q each include a differentialcircuit that generates output at a positive terminal and output at anegative terminal.

The FLTi (e.g., low-pass filter) removes an unnecessary harmoniccomponent from the output signal from the MIXrx_I. The PGAi amplifies anoutput signal from the FLTi based on the gain corresponding to an inputrange for the ADCi. The ADCi converts an output signal from the PGAiinto a digital signal. The FLTq (e.g., low-pass filter moves anunnecessary harmonic component from the output signal from the MIXrx_Q.The PGAq amplifies an output signal from the FLTq based on the gaincorresponding to an input range for the ADCq. The ADCq converts anoutput signal from the PGAq into a digital signal. The SWi outputs anoutput signal from the ADCi to one of the BE and the CALBK. The SWqoutputs an output signal from the ADCq to one of the BE and the CALBK.

The RFIC illustrated in FIG. 2 includes normal operation mode andcalibration mode. In the normal operation mode, the SWr selects the LNAside as an input source. The SWi and the SWq select the BE side as anoutput destination. As an ordinary reception operation, the receptionpower signal RX from the antenna ANT illustrated in FIG. 1 is convertedinto the baseband frequency band and then is supplied to the BE. Thecalibration mode is used as needed for a period in the normal operationmode (e.g., a power-on sequence). In the calibration mode, the SWrselects the TSGEN side as an input source. The SWi and the SWq selectthe CALBK side as an output destination.

The TSGEN generates a test signal RFtst that results from modulating ahigh-frequency signal having a specified frequency f_tx with a 0.5 MHzsignal, for example. Therefore, the RFtst contains a frequency componentof f_tx±0.5 MHz. The frequency f_tx is set equally to the local signalLOtx illustrated in FIG. 1, for example. The RFtst is input to theMIXrx_I and the MIXrx_Q via the SWr. As described above, the MIXrx_I andthe MIXrx_Q perform downconversion using the LOrx_I and the LOrx_Q. Thesecondary intermodulation distortion (IM2) occurs if a differentialbalance difference exits in each of the MIXrx_I and the MIXrx_Q. The IM2frequency component is one MHz in this example. The ADCi and the ADCqconvert the 1-MHz IM2 frequency component into digital signals that arethen input to the CALBK via the SWi and the SWq.

The correction circuit block CALBK includes amplifier circuits LAMPi andLAMPq, a phase detection circuit PHDET, and a digital correction circuitDGCTL. The LAMPi amplifies an output signal (1-MHz in this example) fromthe SWi. The LAMPq amplifies an output signal (1-MHz in this example)from the SWq. The PHDET detects a phase of an output signal from theLAMPi and a phase of an output signal from the LAMPq. The DGCTL changesthe differential balance for the MIXrx_I according to a result ofdetecting the phase of the output signal from the LAMPi in the PHDET.The DGCTL also changes the differential balance for the MIXrx_Qaccording to a result of detecting the phase of the output signal fromthe LAMPq in the PHDET.

The CALBK searches for a differential balance equivalent to a transitionpoint at which the phase reverses approximately at 180° in the outputsignal from the LAMPi, while allowing the DGCTL to appropriately changethe differential balance for the MIXrx_I. Similarly, the CALBK searchesfor a differential balance equivalent to a transition point at which thephase reverses approximately at 180° in the output signal from theLAMPq, while allowing the DGCTL to appropriately change the differentialbalance for the MIXrx_Q. In the normal operation mode, the MIXrx_I andthe MIXrx_Q operate using the differential balance as a search result ofthe CALBK.

As described above, the phase for the IM2 component reverses at 180°corresponding to the differential balance that minimizes the IM2component magnitude. The high frequency signal processor according tothe first embodiment uses this characteristic to detect the 180°reversal of the phase and correct the differential balance. This caneasily search for and highly accurately detect the minimum value for theIM2 component. As described above, the technique of detecting amplitudelevels for the IM2 component requires comparing fine magnituderelationship among amplitude levels for the IM2 component. The detectionoperation itself may be difficult. Highly accurate detection may bedifficult. The amplifier circuit (e.g., LAMPi or LAMPq in FIG. 2) maypreviously amplify the amplitude level to facilitate the detection orimprove the detection accuracy. However, the amplifier circuit requiresstrict linearity because the technique compares the magnituderelationship among amplitude levels. As a result, the gain is limited.

The phase detection technique according to the first embodiment cansolve this problem because the technique just needs to detect an obviouschange, namely, the 180° reversal of the phase. Even if the IM2component contains minute amplitude levels, the LAMPi and the LAMPqillustrated in FIG. 2 can amplify the amplitude levels to sufficientlevels and detect the phase. The LAMPi and the LAMPq do not requirelinearity. The technique can use a high-gain amplifier circuit such as alimiting amplifier.

FIG. 3A is an explanatory diagram illustrating an example of IM2characteristics in the high-frequency signal processor illustrated inFIG. 2. FIG. 3B is a conceptual diagram illustrating an example ofcharacteristic mechanism illustrated in FIG. 3A. FIG. 3A at F301 variesan IM2 correction parameter (e.g., the differential balance for theMIXrx_I in FIG. 2). A specific correction parameter minimizes the IM2and provides an optimum correction point (an optimum differentialbalance for the MIXrx_I in FIG. 2). As illustrated in FIG. 3A at F302,the phase drastically reverses 180° at the optimum correction point. Thecorrection circuit block CALBK in FIG. 2 searches for an optimumcorrection point using the phase characteristic indicated by F302.

FIG. 3B conceptually illustrates IM2 components using vectors. In FIG.3B, vector I represents an IM2 component the mixer circuit MIXrx_I (orMIXrx_Q) in FIG. 2 generated on the positive terminal side. Vector IBrepresents an IM2 component the mixer circuit MIXrx_I (or MIXrx_Q) inFIG. 2 generated on the negative terminal side. Initially, vectors I andIB belong to different phases as illustrated in FIG. 3B at F303. Asdifferential output proceeds, resultant IM2 vector IM2 for a first phaseoccurs and corresponds to vector I-IB. In this state, varying the IM2correction parameter counterclockwise rotates the phase for vector I andclockwise rotates the phase for vector IB. Vectors I and IB approximateto each other. As a result, the resultant IM2 vector IM2 keeps its phaseunchanged and decreases its magnitude.

Vectors I and IB illustrated in FIG. 3B ideally result in I=IB at theoptimum correction point illustrated in FIG. 3A. As a result, theresultant IM2 vector IM2 (1-IB) equals to zero. Then, further varyingthe IM2 correction parameter counterclockwise rotates the phase forvector I and clockwise rotates the phase for vector IB. As a result,vectors I and IB separate from each other as illustrated in FIG. 3B atF304. At this stage, the resultant IM2 vector IM2 (I-IB) has a secondphase different from the first phase 180° at F303. As the IM2 correctionparameter varies, the resultant IM2 vector IM2 increases its magnitudewhile maintaining the second phase. This example assumes that vectors Iand IB have the same magnitude. Actually, vectors I and IB may haveslightly different magnitudes. Also in this case, the phase of theresultant IM2 vector IM2 changes approximately 180° at the optimumcorrection point.

Details of the Test Signal Generating Circuit and the Correction CircuitBlock

FIG. 4 is a circuit block diagram illustrating a more detailedconfiguration example of a test signal generating circuit and acorrection circuit block in the high-frequency signal processorillustrated in FIG. 2. FIG. 4 illustrates a detailed configurationexample of the test signal generating circuit TSGEN and the correctioncircuit block CALBK illustrated in FIG. 2. The LNA, the MIXrx_I, theMIXrx_Q, the LOGEN, the FLTi, the FLTq, the PGAi, and the PGAq areconfigured to be differential. The ADCi and the ADCq receivedifferential output signals from the PGAi and the PGAq, respectively,and convert the signals into a single digital signal.

In FIG. 4, the TSGEN includes a test local signal generating circuitLOTSG, a test baseband signal generating circuit BBTSG, a dividercircuit DIVN, and a test mixer circuit MIXtst. The LOTSG generates atest carrier signal having a specified frequency f_tx. The frequencyf_tx is favorably set to be equal to the frequency of the local signalLOtx for transmission illustrated in FIG. 1. In this case, the LOTSG maybe used in common with a local signal generating circuit (notillustrated) for transmission that generates the LOtx. This can preventa circuit area from increasing.

The BBTSG generates an oscillation signal having a specified frequency(e.g., one MHz) corresponding to the baseband band. The DIVN divides anoscillation signal from the BBTSG by a specified division ratio (e.g.,2). The MIXtst multiplies a carrier signal from the LOTSG by an outputsignal (e.g., 0.5-MHz oscillation signal) from the DIVN. In other words,a carrier signal from the LOTSG up-converts an output signal from theDIVN. As a result, the MIXtst generates the test signal RFtst having afrequency component of f_tx±0.5 MHz. In this example, the LOTSG, theBBTSG, the DIVN, and the MIXtst are configured to be differential.

As illustrated in FIG. 2, the RFtst is input to the mixer circuitsMIXrx_I and MIXrx_Q via the switch SWr. The ADCi and the ADCq convertoutput signals from the MIXrx_I and the MIXrx_Q into digital signals.Digital signals from the ADCi and the ADCq are input to the correctioncircuit block CALBK via switches SWi and SWq, respectively. The CALBKincludes bandpass filters BPFi and BPFq, amplifier circuits LAMPi andLAMPq, a phase detection circuit PHDET, a digital correction circuitDGCTL, and an analog-digital converter circuit ADCtst. The DGCTLprovides a logical operation circuit that performs digital processing.The DGCTL can represent but is not limited to a state machine or asmall-scale processor, for example.

The BPFi extracts an IM2 component (e.g., 1-MHz component) from thedigital signal supplied from the ADCi via the SWi while the IM2component is generated from the MIXrx_I. The BPFq extracts an IM2component (e.g., 1-MHz component) from the digital signal supplied fromthe ADCq via the SWq while the IM2 component is generated from theMIXrx_Q. The LAMPi amplifies an output signal from the BPFi to asufficient level. The LAMPq amplifies an output signal from the BPFq toa sufficient level. In this example, the BPFi and the BPFq providedigital filters. The LAMPi and the LAMPq provide digital amplifiers. TheADCtst converts an oscillation signal (e.g., one MHz) from the BBTSGinto a digital signal. The ADCtst generates a test reference oscillationsignal REFtst as a digital signal.

The PHDET detects a phase for the output signal from the LAMPi and aphase for the output signal from the LAMPq with reference to the phasefor the REFtst (e.g., one MHz). The DGCTL changes the differentialbalance for the MIXrx_I according to a phase detection result for theLAMPi from the PHDET. The DGCTL changes the differential balance for theMIXrx_Q according to a phase detection result for the LAMPq from thePHDET. Specifically, the DGCTL sets a differential balance (assumed tobe setting [1]) for the MIXrx_I. The DGCTL acquires a phase detectionresult (assumed to be result [1]) for the LAMPi, namely, a phasedifference between the phase for the LAMPi and the phase for the REFtst.The DGCTL then appropriately changes the differential balance (assumedto be setting [2]) for the MIXrx_I. The DGCTL acquires a phase detectionresult (assumed to be result [2]) for the LAMPi, namely, a phasedifference between the phase for the LAMPi and the phase for the REFtst.

The DGCTL finds a phase difference between the above-mentioned results[1] and [2]. In FIG. 3A at F302, suppose that a minimum fluctuationrange results from the differential balance (IM2 correction parameter)between the above-mentioned results [1] and [2]. If the phase differencebetween the results [1] and [2] is approximately 180°, the setting [1]or [2] provides the optimum correction point. If the phase differencebetween the results [1] and [2] is approximately 0°, the DGCTL assumesthat another differential balance may contain the optimum correctionpoint. The DGCTL then changes the differential balance. The DGCTLrepeats this process to search for the optimum correction point for theMIXrx_I. After acquiring the optimum correction point for the MIXrx_I,the DGCTL similarly searches for the optimum correction point for theMIXrx_Q.

There has been described a phase difference of approximately 180° or 0°between the differential balance (IM2 correction parameter) and theminimum fluctuation range. Actually, however, some errors may occur. Ifthe minimum fluctuation range is very small, for example, the optimumcorrection point may cause a phase difference somewhat smaller thanapproximately 180°. The DGCTL can actually use 90° as a determinationthreshold value, though not limited thereto. The DGCTL determines thatthe optimum correction point is available if the phase difference fromthe minimum fluctuation range is larger than or equal to 90°. The DGCTLdetermines that another differential balance may contain the optimumcorrection point if the phase difference is smaller than 90°. Theinvention is not limited to this technique. For example, a possibletechnique may use more than one determination threshold value and searchfor a position corresponding to the largest phase difference. Whichevertechnique is used, the phase difference from the minimum fluctuationrange obviously decreases (ideally 0°) if the differential balancedeviates from the optimum correction point. This characteristic can beused to estimate the optimum correction point early. The phasedifference obviously increases (ideally 180°) at the optimum correctionpoint. This characteristic can be used to estimate the optimumcorrection point highly accurately.

As illustrated in FIGS. 4 and 2, input to the CALBK follows the ADCi andthe ADCq. The invention is not limited thereto. Input to the CALBK justneeds to follow the MIXrx_I and the MIXrx_Q. For example, output fromthe PGAi and the PGAq may be input to the CALBK. After the ADCi and theADCq, the CALBK performs digital processing for amplification and phasedetection. After the PGAi and the PGAq, the CALBK performs analogprocessing for amplification and phase detection. As described above,however, the CALBK compares the phase (result [2]) at a given time pointwith the phase (result [1]) at another time point. The digitalprocessing is considered more appropriate than the analog processing.Input to the CALBK favorably follows the ADCi and the ADCq from theviewpoint of a noise that may result from the switches SWi and SWq. TheSWr may be positioned before the MIXrx_I and the MIXrx_Q and favorablyafter the LNA from the viewpoint of a noise or a noise figure (NF).

Details of the Reception Mixer Circuit

FIG. 5 is a circuit diagram illustrating a configuration example of areception mixer circuit in the high-frequency signal processorillustrated in FIG. 2. A mixer circuit MIXrx illustrated in FIG. 5 isequivalent to each of the MIXrx_I and the MIXrx_Q illustrated in FIG. 2.The MIXrx includes two pairs of differential pair transistors MNDP1 andMNDP2, a phase shift circuit PHSFT, and a back gate (substratepotential) control circuit BGCTL. FIG. 5 also illustrates the localsignal generating circuit LOGEN illustrated in FIG. 2. The PHSFTgenerates a signal by adding a specified phase difference (e.g., 180°)to the local signal from the LOGEN. The MNDP1 includes two NMOStransistors MN1 a and MN1 b whose sources are coupled to each other incommon. The source is supplied with a high-frequency signal RFin fromthe positive terminal. The MNDP2 includes two NMOS transistors MN2 a andMN2 b whose sources are coupled to each other in common. The source issupplied with a high-frequency signal (/RFin) from the negativeterminal. The RFin and the /RFin are equivalent to output signals fromthe SWr in FIG. 2.

The gates of the MN1 a and the MN2 b are supplied with a local signalfrom the LOGEN. The gates of the MN1 b and the MN2 a are supplied with alocal signal from the LOGEN via the PHSFT. The drain of the MN1 a iscoupled to the drain of the MN2 a in common. The drain generates anoutput signal (current signal) I at the positive terminal. The drain ofthe MN1 b is coupled to the drain of the MN2 b in common. The draingenerates an output signal (current signal) IB at the negative terminal.The BGCTL appropriately controls back bias (substrate potential) for theMN1 a, the MN1 b, the MN2 a, and the MN2 b. The MIXrx in FIG. 5 isreferred to as a passive double-balanced mixer (DBM). The MIXrx_I andthe MIXrx_Q illustrated in FIG. 2 are not limited to the configurationillustrated in FIG. 5 and may use an active DBM such as the Gilbert cellor a single-balanced mixer according to circumstances. However, theconfiguration in FIG. 5 is favorable from the viewpoint of powerconsumption, linearity, and high speed.

FIGS. 6 through 8 are conceptual diagrams illustrating different examplemethods for adjusting the differential balance (IM2 correctionparameter) in the mixer circuit illustrated in FIG. 5. FIGS. 6 through 8conceptually illustrate relationships among the configuration exampleFIG. 5, the phase detection circuit PHDET, and the digital correctioncircuit DGCTL. As illustrated in FIGS. 6 through 8, the differentialoutput signal (I or IB) from the mixer circuit MIXrx contains theresultant IM2 vector that results from a vector difference (I-IB)between the IM2 component of the signal I and the IM2 component of thesignal IB. The PHDET detects the phase for the resultant IM2 vector. TheDGCTL monitors a result of the PHDET that detects the phase for theresultant IM2 vector. The DGCTL appropriately changes the differentialbalance (IM2 correction parameter) while determining whether the phaseis reversed.

According to the example in FIG. 6, the DGCTL changes the differentialbalance (IM2 correction parameter) in the phase shift circuit PHSFT. Forexample, the DGCTL allows the PHSFT to vary the phase shift amount inthe range of approximately 180°. The DGCTL thereby changes thedifferential balance between the MN1 a and the MN1 b the differentialbalance between the MN2 a and the MN2 b. According to an example in FIG.7, the DGCTL allows the back gate control circuit BGCTL to change thedifferential balance. For example, the DGCTL allows the BGCTL to controlthe back gates of the transistors MN1 a, MN1 b, MN2 a, and MN2 b tochange threshold voltages for these transistors. The BGCTL therebychanges the differential balance.

According to an example in FIG. 8, the DGCTL allows a load circuit LOADto change the differential balance. As illustrated in FIG. 8, the mixercircuit MIXrx is actually provided with a load (e.g., a resistorelement) that converts the output signals (current signals) I and IBinto voltage signals. The DGCTL changes the differential balance bychanging the relative balance between the I-side load magnitude and theIB-side load magnitude. As illustrated in FIGS. 6 through 8, varioustechniques can be used to change the differential balance for the MIXrx.The embodiment is not limited to the techniques of changing thedifferential balance and just needs to select the technique according tothe mixer circuit scheme. If the passive double-balanced mixer asillustrated in FIG. 5 is used, for example, the embodiment can use anyone of or a combination of the techniques illustrated in FIGS. 6 through8. If the active double-balanced mixer is used, for example, theembodiment may use a technique of changing the bias current balance foreach of the differential pair transistors.

Searching for an Optimum Correction Point (Binary Search)

FIG. 9 is a flowchart illustrating an example method for the correctioncircuit block in the high-frequency signal processor illustrated in FIG.2 to search for an optimum correction point. FIG. 10 supplements anexample of actual operation according to the flowchart illustrated inFIG. 9. As described above, the correction circuit block CALBK (e.g.,the digital correction circuit DGCTL) illustrated in FIG. 2 searches foran optimum correction point using the characteristic that the phase forthe resultant IM2 vector transitions approximately 180° at the optimumcorrection point. The CALBK can therefore perform the binary search asillustrated in FIGS. 9 and 10.

FIG. 9 assumes m-bit wide control (a variable range of 2^(m) bits) ofthe IM2 correction parameter. The correction circuit block CALBKacquires a reference phase (ST[0]). The reference phase just needs tocorrespond to one of both ends of the variable range. In this example,the reference phase corresponds to the zeroth point as a determinationpoint. The CALBK sets the IM2 correction parameter to point 2^((m-1)),namely, a middle point in the variable range. As the firstdetermination, the CALBK compares the phase at the determination point(2^((m-1))) with the reference phase (ST[1]). If the phase reverses(e.g., transition of approximately 180°) as a comparison result atST[1], the CALBK proceeds to ST[2]a and performs the seconddetermination. If the phase does not reverse (e.g., transition ofapproximately 0°), the CALBK proceeds to ST[2]b and performs the seconddetermination. As described above, the CALBK just needs to determinewhether the phase reverses by checking if a phase difference from thereference phase is larger than or equal to 90° or smaller than 90°,though not limited thereto.

At ST[2]a, the CALBK sets the IM2 correction parameter to adetermination point 2^((m-1))−2^((m-2)), found by subtracting 2^((m-2))from the first determination point, and compares the phase at thedetermination point with the reference phase. At ST[2]b, the CALBK setsthe IM2 correction parameter to a determination point2^((m-1))+2^((m-2)), found by adding 2^((m-2)) to the firstdetermination point, and compares the phase at the determination pointwith the reference phase. If the phase reverses as a result of the firstdetermination, the phase transition point exits between the first pointand point 2^((m-1)). At ST[2]a, the CALBK verifies the phase at themiddle point. If the phase does not reverse, the phase transition pointexits between point 2^((m-1)) and point 2^(m). At ST[2]b, the CALBKverifies the phase at the middle point.

If the phase reverses at ST[2]a or ST[2]b, the CALBK similarly proceedsto ST[3]a (not illustrated) and performs the third determination. If thephase does not reverse, the CALBK similarly proceeds to ST[3]b (notillustrated) and performs the third determination. Similar processesfollow. The nth determination includes ST[n]a and ST[n]b. At ST[n]a, theCALBK compares the reference phase with the phase at a determinationpoint found by subtracting 2^((m-1)) from the (n−1)th determinationpoint. At ST[n]b, the CALBK compares the reference phase with the phaseat a determination point found by adding 2^((m-1)) to the (n−1)thdetermination point. If the phase reverses at ST[n]a or ST[n]b, theCALBK proceeds to ST[n+1]a (not illustrated) and performs the (n+1)thdetermination. If the phase does not reverse, the CALBK proceeds toST[n]b (not illustrated) and performs the (n+1)th determination.

Finally, the mth determination includes ST[m]a and ST[m]b. At ST[m]a,the CALBK compares the reference phase with the phase at a determinationpoint found by subtracting 2^((m-m)) from the (m−1)th determinationpoint. At ST[m]b, the CALBK compares the reference phase with the phaseat a determination point found by adding 2^((m-m)) to the (m−1)thdetermination point. If the phase reverses at ST[m]a or ST[m]b, theCALBK proceeds to ST[m+1]a. If the phase does not reverse, the CALBKproceeds to ST[m+1]b. At ST[m+1]a, the CALBK registers the optimumcorrection point corresponding to the determination point at the mthdetermination. At ST[m+1]b, the CALBK registers the optimum correctionpoint corresponding to a point found by adding 1 to the determinationpoint at the mth determination.

An example in FIG. 10 assumes that the IM2 correction parameter isapplicable to a variable range of m=6 (2^(m)=64 points) and that the52nd point corresponds to the optimum correction point. At the firstdetermination, the reference phase (phase at the zeroth point) isreverse to the phase at the 32nd point. The CALBK determines that theoptimum correction point exists between the 32nd point and the 64thpoint. The CALBK performs the second determination, assuming the 48thpoint as the middle point to be a determination point. At the seconddetermination, the reference phase is not reverse to the phase at the48th point. The CALBK determines that the optimum correction pointexists between the 48th point and the 64th point. The CALBK performs thethird determination, assuming the 56th point as the middle point to be adetermination point.

At the third determination, the reference phase is reverse to the phaseat the 56th point. The CALBK determines that the optimum correctionpoint exists between the 48th point and the 56th point. The CALBKperforms the fourth determination, assuming the 52nd point as the middlepoint to be a determination point. At the fourth determination, thereference phase is reverse to the phase at the 52nd point. The CALBKdetermines that the optimum correction point exists between the 48thpoint and the 52nd point. The CALBK performs the fifth determination,assuming the 50th point as the middle point to be a determination point.At the fifth determination, the reference phase is not reverse to thephase at the 50th point. The CALBK determines that the optimumcorrection point exists between the 50th point and the 52nd point. TheCALBK performs the sixth determination, assuming the 51st point as themiddle point to be a determination point. At the sixth determination,the reference phase is not reverse to the phase at the 51st point. As aresult, the CALBK determines the 52nd point to be the optimum correctionpoint.

According to the flowchart illustrated in FIG. 9, the CALBK divides thevariable range of 2^(m) bits for the IM2 correction parameter in half.The CALBK then determines which part of the range contains the optimumcorrection point (phase transition point). The CALBK further divides therange containing the optimum correction point in half and determineswhich part of the range contains the optimum correction point. Thisprocess is performed m times. The determination process, when performedm times, narrows the range for the 1-bit optimum correction point. Thebinary search just needs to perform the determination process m timesfor the search range of 2^(m) bits. The optimum correction point can befound in a short search time.

As a comparison, consider a case of searching for the optimum correctionpoint using the amplitude level detection as illustrated in FIG. 3A atF301. In this case, it is difficult to use the binary search. Forexample, the following search technique needs to be used. Suppose thevariable range is 64 bits. For example, the search technique divides therange by eight bits to yield eight determination points. The searchtechnique compares IM2 amplitude levels at the determination points tosearch for a determination point corresponding to the smallestamplitude. The search technique sweeps a range of ±4 bits one bit by oneat the determination point to search for the minimum point for theamplitude. This search technique requires 16 determination processes intotal. By contrast, the search technique illustrated in FIG. 9 requiresonly six determination processes in total.

The high-frequency signal processor and the wireless communicationsystem according to the first embodiment of the invention can typicallyallow the correction circuit block to perform a correction process andreduce the secondary intermodulation distortion that may occur in thereception circuit. The correction circuit block detects the phasetransition point of an IM2 vector to search for the optimum correctionpoint. This enables to facilitate the search operation, increase thesearch accuracy, and shorten the search time.

Second Embodiment

FIGS. 11A and 11B illustrate an example problem in a high-frequencysignal processor according to a second embodiment of the invention. FIG.11A illustrates a high-frequency signal processor RFIC having theconfiguration similar to that illustrated in FIG. 2. The firstembodiment provides the example that independently corrects the I-side(mixer circuit MIXrx_I) and the Q-side (mixer circuit MIXrx_Q) in FIG.11A. Actually, however, the MIXrx_I and the MIXrx_Q may operatesimultaneously. As illustrated in FIG. 11A, an IQ interference may occurdue to a leak signal LK_IM2 containing an IM2 component between theMIXrx_I and the MIXrx_Q.

If the IM2 correction parameter is varied for the I-side (MIXrx_I), theoptimum correction point occurs on the Q-side (MIXrx_Q) as well as theI-side as illustrated in FIG. 11B. This situation may occur even if theI-side and the Q-side are reversed. For example, an IM2 componentresulting from the MIXrx_I leaks to the MIXrx_Q and is superposed on anIM2 component resulting from the MIXrx_Q. This situation may occur evenif the I-side and the Q-side are reversed. If the IQ interferenceoccurs, a correction parameter for the I-side affects the Q-side and acorrection parameter for the Q-side affects the I-side. Just correctingthe I-side and the Q-side independently may not provide the optimumcorrection point for actual operation. The search time may increase ifthe I-side correction and the Q-side correction are alternately repeatedin consideration of the IQ interference. To solve this problem, thesecond embodiment provides a technique of searching for the optimumcorrection point in consideration of influence of the IQ interference.

FIG. 12 is an explanatory diagram illustrating an example method for thehigh-frequency signal processor according to the second embodiment ofthe invention to search for an optimum correction point. At S1201, thecorrection circuit block CALBK in FIG. 11A varies an IM2 correctionparameter (Pi) for the I-side and searches for an optimum correctionpoint (Ii) that minimizes the value of an IM2 component (IM2_I) for theI-side. The IQ interference changes an IM2 component (IM2_Q) for theQ-side and yields a point (Qi) that minimizes the IM2 Q. The CALBKsearches for Qi as well as Ii. At S1201, an IM2 correction parameter(Pq) for the Q-side is unchanged and is fixed to the default (0).

At S1202, the CALBK varies an IM2 correction parameter (Pq) for theQ-side and searches for an optimum correction point (Qq) that minimizesthe value of an IM2 component (IM2_Q) for the Q-side. The IQinterference changes an IM2 component (IM2_I) for the I-side and yieldsa point (Iq) that minimizes the IM2_I. The CALBK searches for Iq as wellas Qq. At S1202, an IM2 correction parameter (Pi) for the I-side isunchanged and is fixed to the default (0). FIG. 12 illustrates the IM2_Iand the IM2_Q using amplitude levels for convenience sake. An actualsearch method uses the phase information as described in the firstembodiment.

Finally, at S1203, the CALBK uses Ii, Qi, Iq, and Qq found at S1201 andS1202 to perform equations (1) and (2) as follows. Equation (1)calculates an optimum correction point Ical for the I-side inconsideration of the IQ interference. Equation (2) calculates an optimumcorrection point Qcal for the Q-side in consideration of the IQinterference.Ical=Ii·Qi·(Iq−Qq)/(Iq·Qi−Ii·Qq)  (1)Qcal=Qq·Iq·(Qi−Ii)/(Qi·Iq−Qq·Ii)  (2)

Equations (1) and (2) are derived as follows. FIGS. 13A and 13Bsupplement FIG. 12. As illustrated in FIGS. 13A and 13B, the verticalaxis represents IM2_I [V]. The horizontal axis represents the IM2correction parameter (Pi) (FIG. 13A) for the I-side and the IM2correction parameter (Pq) (FIG. 13B) for the Q-side. The dependencebetween them is assumed linear. The Ical represents a Pi valuecorresponding to the minimum IM2_I and the Qcal represents a Pq valuecorresponding to the minimum IM2_Q after considering the IQinterference. The Ical and the Qcal are assumed to have the relationshipexpressed in equation (3) below.

$\begin{matrix}{{Ical} = {{Ii} - {\frac{\partial{Pi}}{\partial{Pq}}{Qcal}}}} & (3)\end{matrix}$

Equation (4) derives from the geometric relationship between FIGS. 13Aand 13B. Reflecting equation (4) on equation (3) derives equation (5).Equation (6) is similarly derived for the Q-side. Equations (5) and (6)are solved as simultaneous equations to derive equations (1) and (2)described above.

$\begin{matrix}{{\frac{{\partial{IM}}\; 2{\_ I}}{\partial{Pq}}\text{:}\frac{{\partial{IM}}\; 2{\_ I}}{\partial{Pi}}} = {{Ii}\text{:}{Iq}}} & (4) \\{{Ical} = {{{Ii} - {{\frac{\partial{Pi}}{{\partial{IM}}\; 2{\_ I}} \cdot \frac{{\partial{IM}}\; 2{\_ I}}{\partial{Pq}}}{Qcal}}} = {{Ii} - {\frac{Ii}{Iq} \cdot {Qcal}}}}} & (5) \\{{Qcal} = {{Qq} - {\frac{Qq}{Qi} \cdot {Ical}}}} & (6)\end{matrix}$

The high-frequency signal processor according to the second embodimentcan search for the optimum correction point more highly accurately inconsideration of the influence of the IQ interference in addition tovarious effects described in the first embodiment. Particularly, thepoint Qi or Iq in FIG. 12 may decrease the amplitude level. The point Qior Iq may be detected with difficulty at the amplitude level. However,the embodiment detects these points using phases. The amplifier circuitsLAMPi and LAMPq in FIG. 11A can sufficiently amplify the points andeasily and highly accurately detect the points. The search time can beshortened because the search for Ii, Qi, Iq, and Qq just requires 24determination processes according to the example in FIG. 10 (m=6). Bycontrast, more than 24 determination processes may be required until thefinal detection if the correction of the I-side and the Q-side isalternately repeated several times in consideration of the IQinterference.

Third Embodiment

The third embodiment describes an optimum correction point searchtechnique different from the second embodiment. FIG. 14 is a flowchartillustrating an example method for a high-frequency signal processoraccording to the third embodiment of the invention to search for anoptimum correction point. The high-frequency signal processor accordingto the third embodiment is configured similarly to the high-frequencysignal processor RFIC as illustrated in FIGS. 2 and 4. At S1401 in FIG.14, the high-frequency signal processor RFIC turns on or activates thereception frontend block (the reception circuits in FIG. 1 such as LNA,MIXrx, and BE or the circuits in FIG. 2 other than TSGEN). At S1402, theRFIC then turns on or activates the test signal generating circuit(TSGEN in FIGS. 2 and 4).

At S1403, the RFIC (e.g., the correction circuit block CALBK) turns offor inactivates the Q-side mixer circuit (MIXrx_Q in FIGS. 2 and 4). Inthis state, the RFIC corrects the I-side IM2 or searches for the IM2correction parameter for the MIXrx_I in FIGS. 2 and 4 using the testsignal generating circuit TSGEN and the CALBK. At S1404, the RFIC (e.g.,the CALBK) turns off the I-side mixer circuit (the MIXrx_I in FIGS. 2and 4) and turns on the Q-side mixer circuit. In this state, the RFICcorrects the Q-side IM2 or searches for the IM2 correction parameter forthe MIXrx_Q in FIGS. 2 and 4 using the TSGEN and the CALBK. The mixercircuit is turned off or inactivated if input of the local signal,LOrx_I or LOrx_Q, stops or the buffer circuit to output LOrx_I or LOrx_Qstops, for example. However, the invention is not limited thereto.

The IM2 correction parameters for the MIXrx_I and the MIXrx_Q aresearched at S1403 and S1404. At S1405, the RFIC registers these IM2correction parameters to the MIXrx_I and the MIXrx_Q and completes theIM2 correction. This search method can correct the IM2 easily and fastwithout being influenced by the IQ interference. Compared to thetechnique according to the second embodiment, the technique according tothe third embodiment may cause an error at the optimum correction pointif the MIXrx_I and the MIXrx_Q operate simultaneously. In this case, thetechnique according to the second embodiment is more favorable. However,the technique according to the second embodiment may cause Qi or Iq inFIG. 12 to exceed the variable range of IM2. In this case, the techniqueaccording to the third embodiment is more favorable.

While there have been described specific preferred embodiments of thepresent invention, it is to be distinctly understood that the presentinvention is not limited thereto but may be otherwise variously embodiedwithin the spirit and scope of the invention.

Particularly, the high-frequency signal processor and the wirelesscommunication system according to the embodiment are beneficiallyapplicable to a high-frequency signal processor having direct conversionreception circuit and performing FDD-based transmission and receptionand applicable to a mobile telephone having the high-frequency signalprocessor. In addition, the high-frequency signal processor and thewireless communication system according to the embodiment are applicableto a high-frequency signal processor performing TDD-based transmissionand reception and applicable to various wireless communication systemssuch as wireless LAN (Local Area Network) and Bluetooth (registeredtrademark).

What is claimed is:
 1. A high-frequency signal processor provided with afirst operation mode and a second operation mode, the high-frequencysignal processor comprising: a test signal generating circuit thatgenerates a test signal having a first frequency component and a secondfrequency component; a first switch that transmits a signal received asa first signal at an antenna in the first operation mode and transmitsthe test signal as the first signal in the second operation mode; amixer circuit that includes a differential circuit capable of correctinga differential balance within a specified variable range anddown-converts the first signal to a second signal having a frequencyband lower than the first signal; a phase detection portion thatextracts a third signal from the second signal in the second operationmode, the third signal having a frequency component equivalent to adifference between the first frequency component and the secondfrequency component, and detects a phase for the third signal; a controlportion that changes the differential balance for the mixer circuitaccording to a detection result from the phase detection portion,wherein the mixer circuit operates in the first operation mode while thedifferential balance is set to a first correction value within avariable range, and wherein the control portion, in the second operationmode, varies the differential balance and concurrently searches for atransition point allowing a phase for the third signal to transition byapproximately 180° before and after varying the differential balancewithin a minimum fluctuation range and supplies the mixer circuit withthe first correction value, namely, the differential balancecorresponding to the transition point; an analog-digital convertercircuit provided after the mixer circuit; a baseband circuit thatperforms a specified baseband process: and a second switch that selectsone out of transmitting an output from the analog-digital convertercircuit to the baseband circuit and transmitting the output from theanalog-digital converter circuit to the phase detection portion, whereinthe phase detection portion receives the second signal as a digitalsignal via the second switch, wherein the test signal generating circuitincludes: a test local signal generating circuit that generates a testlocal signal having a specified frequency; a test baseband signalgenerating circuit that generates a test baseband signal having afrequency equivalent to a difference between the first frequencycomponent and the second frequency component; a divider circuit thatdivides the test baseband signal into two; and a test mixer circuit thatup-coverts an output signal from the divider circuit using the testlocal signal, and wherein the phase detection portion includes: adigital filter circuit that extracts the third signal; a digitalamplifier circuit that amplifies an output signal from the digitalfilter circuit; a test analog-digital converter circuit that convertsthe test baseband signal into a digital signal: and a phase detectioncircuit that detects a phase for an output signal from the digitalamplifier circuit with reference to a phase for an output signal fromthe test analog-digital converter circuit.
 2. The high-frequency signalprocessor according to claim 1, wherein the control portion performs abinary search to sequentially halve a variable range of the differentialbalance and concurrently searches for the differential balancecorresponding to the transition point.
 3. The high-frequency signalprocessor according to claim 1, further comprising: a low-noiseamplifier circuit that is provided along a path from the antenna to thefirst switch.